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LT1614

Inverting 600kHz Switching Regulator

Linear Technology 

LT1614

U

OPERATIO

The LT1614 combines a current mode, fixed frequency

PWM architecture with a –1.23V reference to directly

regulate negative outputs. Operation can be best under-

stood by referring to the block diagram of Figure 2. Q1 and

Q2 form a bandgap reference core whose loop is closed

around the output of the converter. The driven reference

point is the lower end of resistor R4, which normally sits

at a voltage of –1.23V. As the load current changes, the

NFB pin voltage also changes slightly, driving the output

of gm amplifier A1. Switch current is regulated directly on

a cycle-to-cycle basis by A1’s output. The flip-flop is set at

the beginning of each cycle, turning on the switch. When

the summation of a signal representing switch current and

a ramp generator (introduced to avoid subharmonic oscil-

lations at duty factors greater than 50%) exceeds the VC

signal, comparator A2 changes stage, resetting the flip-

flop and turning off the switch. Output voltage decreases

(the magnitude increases) as switch current is increased.

The output, attenuated by external resistor divider R1 and

R2, appears at the NFB pin, closing the overall loop.

Frequency compensation is provided externally by a series

RC connected from the VC pin to ground. Typical values

are 100k and 1nF. Transient response can be tailored by

adjustment of these values.

As load current is decreased, the switch turns on for a

shorter period each cycle. If the load current is further

decreased, the converter will skip cycles to maintain

output voltage regulation.

VIN

+

SHUTDOWN

L1

C1

VIN

SW

LT1614

SHDN

VC

10Ok

NFB

GND

1nF

C2

1µF

R1

R2

10k

D2

D1

–VOUT

C3

The LT1614 can work in either of two topologies. The

simpler topology appends a capacitive level shift to a

boost converter, generating a negative output voltage,

which is directly regulated. The circuit schematic is de-

tailed in Figure 3. Only one inductor is required, and the

two diodes can be in a single SOT-23 package. Output

noise is the same as in a boost converter, because current

is delivered to the output only during the time when the

LT1614’s internal switch is on.

If D2 is replaced by an inductor, as shown in Figure 4, a

higher performance solution results. This converter topol-

ogy was developed by Professor S. Cuk of the California

Institute of Technology in the 1970s. A low ripple voltage

results with this topology due to inductor L2 in series with

the output. Abrupt changes in output capacitor current are

eliminated because the output inductor delivers current to

the output during both the off-time and the on-time of the

LT1614 switch. With proper layout and high quality output

capacitors, output ripple can be as low as 1mVP–P.

The operation of Cuk’s topology is shown in Figures 5

and␣ 6. During the first switching phase, the LT1614’s

switch, represented by Q1, is on. There are two current

loops in operation. The first loop begins at input capacitor

C1, flows through L1, Q1 and back to C1. The second loop

flows from output capacitor C3, through L2, C2, Q1 and

back to C3. The output current from RLOAD is supplied by

L2 and C3. The voltage at node SW is VCESAT and at node

SWX the voltage is –(VIN + |VOUT|). Q1 must conduct both

L1 and L2 current. C2 functions as a voltage level shifter,

with an approximately constant voltage of (VIN + |VOUT|)

across it.

C2

L1

1µF

L2

VIN

+

SHUTDOWN

VIN

SW

C1

LT1614

SHDN

VC

10Ok

NFB

GND

1nF

D1

R1

R2

10k

–VOUT

C3

1614 F03

Figure 3. Direct Regulation of Negative Output

Using Boost Converter with Charge Pump

1614 F04

Figure 4. L2 Replaces D2 to Make Low Output Ripple

Inverting Topology. Coupled or Uncoupled Inductors Can

Be Used. Follow Phasing If Coupled for Best Results

6

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